Described below is a method for coding data symbols which are transmitted in an OFDM radio transmission via at least two transmitting antennas.
In future broadband radio communication systems, multi-carrier transmission methods will be used for transmissions with high data rates. The use of the familiar OFDM—orthogonal frequency division multiplex—transmission method in combination with multi-antenna systems, particularly with MIMO antenna systems, is particularly suitable.
Due to the multipath propagation of radio waves and due to time-dependent changes in the mobile radio channel characteristics, radio signals are distorted during the transmission. To be able to receive transmitted data free of distortion, it is known to use a so-called coherent detection at the receiver end. For this purpose, a radio channel used is measured at regular intervals with the aid of transmitted training sequences or pilot symbols which are previously known at the receiving end. By this, the receiver can determine current distortions of the radio channel via a channel estimation and equalize transmitted data again.
However, such a channel estimation has the following disadvantages:                the transmission of the training sequences or pilot symbols, respectively, causes a considerable signaling overhead,        during the transmission of the training sequences or pilot symbols, respectively, no (payload) data can be transmitted, and        a complex, time-consuming algorithm must be used for channel estimation at the receiving end.        
In a MIMO radio transmission system, the radio channel estimation can be carried out only with great time expenditure due to the large number of antennas since the number of necessary channel estimations is the result of the product of the number of antennas at the transmitting end and of the number of antennas at the receiving end.
In the case of fast changes in the environment which are experienced, for example, by a fast-moving mobile radio communication terminal, the channel conditions or channel characteristics also change rapidly. In this context, a so-called “coherence time”, which is used as a measure of a time variance of the radio channel characteristics, is essentially defined by the speed of the mobile radio communication terminal.
Time intervals used for radio channel estimation must be clearly below the coherence time in order to achieve a current base for subsequent radio channel estimations. It generally holds true that at a higher speed of the terminal, correspondingly more frequent radio channel estimations must be carried out. Correspondingly frequently, training symbols must be sent instead of (payload) data which, in turn, leads to a decrease in the efficiency of the radio transmission system.
To avoid the radio channel estimation, the so-called “Differential Space Time Block Code, DSTBC”, which is described, for example, in “A Differential Detection Scheme for Transmit Diversity”, Tarokh, Jafarkhani, IEEE Journal on Selected Areas in Communications, Volume 18, No. 7, July 2000, has been defined for MIMO radio transmission systems.
In this context, information or data about a difference between a current signal and a previous signal are modulated. It is assumed that the radio channel essentially does not change within the time interval between two successive data symbols, i.e. the coherence time is greater by a multiple than the symbol duration. In this case, the two successive transmitted data symbols are subjected to approximately the same distortion (considered from an arbitrary time t0).
In other words, the difference between the two successive transmitted symbols is influenced only insignificantly by the distortion of the radio channel. The transmitted information can thus be recovered without additional channel equalization.
The differential coding for a MIMO radio communication system with two transmitting antennas can be described, for example, by the following matrix multiplication:
            S      k        =                            S                      k            -            1                          ·                              C            k                    ⁢                                          [                                                                      s                                                            2                      ⁢                                                                                          ⁢                      k                                        +                    1                                                                                                s                                                            2                      ⁢                                                                                          ⁢                      k                                        +                    2                                                                                                                        -                                      s                                                                  2                        ⁢                                                                                                  ⁢                        k                                            +                      2                                        *                                                                                                s                                                            2                      ⁢                                                                                          ⁢                      k                                        +                    1                                    *                                                              ]                    =                        [                                                                      s                                                            2                      ⁢                                                                                          ⁢                      k                                        -                    1                                                                                                s                                      2                    ⁢                    k                                                                                                                        -                                      s                                          2                      ⁢                      k                                        *                                                                                                s                                                            2                      ⁢                      k                                        -                    1                                    *                                                              ]                ·                  [                                                                      c                                                            2                      ⁢                      k                                        +                    1                                                                                                c                                                            2                      ⁢                      k                                        +                    2                                                                                                                        -                                      c                                                                  2                        ⁢                        k                                            +                      2                                        *                                                                                                c                                                            2                      ⁢                      k                                        +                    1                                    *                                                              ]                      ,where the entries of the matrices Sk and Sk-1 contain transmit symbols and the matrix Ck contains information symbols at a time k.
One column of the transmit matrix in each case contains the transmit symbols which are successively sent or transmitted via an antenna. The two row vectors in the matrix Sk are orthogonal to one another which provides for incoherent detection in the receiver. In contrast to the coherent detection, the incoherent detection is not dependent on a radio channel estimation and no training sequences are thus required at the transmitting end.
When two antennas are used at the transmitting end, two previously known symbols must be transmitted correspondingly before the data transmission, for the initialization of the radio transmission and for the decoding at the receiving end, respectively.
FIG. 1 shows a radio transmission with “Differential Space Time Block Code, DSTBC” in an OFDM radio communication system with two antennas at the transmitting end. In this arrangement, symbols s are series/parallel converted, DSTBC-coded, mapped onto N subcarriers and transmitted orthogonally with respect to one another via two transmitting antennas.
FIG. 2 shows, with reference to FIG. 1, the signaling overhead due to the initialization and the procedure in the differential coding (DSTBC) per subcarrier plotted over time. It can be seen that due to the coding by DSTBC a separate initialization is necessary for each subcarrier in the OFDM radio transmission. It can also be seen that, due to two transmitting antennas being used and due to the DSTBC, in each case two symbols are used per subcarrier for the initialization.
According to the 3GPP TR 25.814 standard, “Physical Layer Aspects for Evolved UTRA”, Release 7, page 17, 2005-11, a maximum of seven symbols are located in a subframe of 500 μsec duration, with a subcarrier spacing of 15 kHz. In this case, the subcarrier spacing in an OFDM system is inversely proportional to the duration of an OFDM symbol and each OFDM symbol has a guard interval.
Due to the orthogonal symbol transmission and the DSTBC coding, the transmission of an OFDM symbol considered also determines the transmission of the next OFDM symbol following so that lastly only six of seven symbols are transmitted in a subframe. This is shown by way of example in FIG. 3.
Together with the initialization in a first OFDM symbol, the overhead amounts to a total of 33% with a 500 μsec subframe and a subcarrier spacing of 15 kHz—i.e. only 66% of the frame are used for a (payload) data transmission.
In the case where there is a number of subscribers, data are transmitted in time-division multiplex (e.g. with TDMA) in a frame, one subframe being allocated to each subscriber. This is shown in FIG. 4A.
If, as shown in FIG. 4B, only one common initialization is used at the beginning of a frame in the downlink in order to be able to reduce a total overhead, a subscriber T2 would have to additionally detect the data of a subscriber T1 in order to be able to incoherently detect data intended for him. If power control is used, this may lead to subscriber T2 not receiving the data signal intended for subscriber T1 in sufficient strength.
To make the differential coding more effective for the combination of MIMO antenna system and OFDM radio transmission, a so-called “Differential Space Time Frequency Block Code, DSTFBC” was presented in “Differential Space-Time-Frequency Transmit Diversity in OFDM” by G. Bauch, Proc. of International Symposium on Wireless Personal Multimedia Communications (WPMC), Yokosuka, Japan, October 2003, which can be used in the symbol coding.
At the core, this is coding by DSTBC, but transmit symbols are distributed not exclusively over time but also over frequency.
The entries of a transmit matrix Sk are sent on a subcarrier in two successive OFDM symbols, the entries of the next matrix are sent on the next subcarrier in each case in the same two OFDM symbols.
FIG. 5 shows the principle of coding by DSTFBC, whilst FIG. 6 shows the resultant overhead for the initialization and the procedure in the differential coding DSTFBC over frequency and time.
For the initialization, the coding by DSTFBC only needs two data symbols on one subcarrier which are in each case sent out via both transmitting antennas according to FIG. 5.
This provides for a distinct reduction in overhead. From 256 subcarriers onward, the total overhead is only 0.1% with a subcarrier spacing of 15 kHz and with a temporal frame duration of 500 μsec. Thus, 99.9% of the frame are available for the data transmission. This corresponds to an increase by 49.5% compared with the coding by DSTBC described initially.
However, the coding by DSTFBC has the decisive disadvantage of poorer performance depending on the OFDM parameterization selected and/or the prevailing fluctuation characteristics of the mobile radio channel. In the case of strong frequency selectivity with respect to the subcarrier spacing and little time variance with respect to the symbol duration, the coding is poorer in the frequency direction than the coding over the time axis. The reason for this is that, for successful coding by DSTFBC, a mobile radio channel should be present which is constant within certain limits.
Although the characteristics of the mobile radio channel change slowly in time, the transfer function of the mobile radio channel exhibits a distinct frequency selectivity in the frequency domain which is due to the multipath propagation. The frequency selectivity is expressed in short, but very deep dips or nulls. At these points, the similarity of the channel of adjacent subcarriers required for the coding by DSTFBC in the frequency domain is given only very inadequately. This results in high bit error rates in the case of coding by DSTFBC in comparison with coding by DSTBC.
Independently of the OFDM parameterization, the abovementioned channel characteristics occur, for example, typically in so-called broadband “fixed wireless access” systems in which, due to the wide bandwidth of the channel, a correspondingly high frequency selectivity can be expected and the transmitting and receiving stations do not move, or only from time to time.